Design of a radiofrequency (RF) energy harvesting system for low-power sensor applications at microwave bands

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(1)UNIVERSIDAD POLITÉCNICA DE MADRID ESCUELA TÉCNICA SUPERIOR DE INGENIEROS DE TELECOMUNICACIÓN. MASTER UNIVERSITARIO EN INGENIERÍA DE TELECOMUNICACIÓN. TRABAJO FIN DE MASTER. TITULO: DESIGN OF A RADIOFREQUENCY (RF) ENERGY HARVESTING SYSTEM FOR LOW-POWER SENSOR APPLICATIONS AT MICROWAVE BANDS. AUTOR: Antonio Alex Amor. AÑO: 2018.

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(3) TÍTULO:. DESIGN OF A RADIOFREQUENCY (RF) ENERGY HARVESTING SYSTEM FOR LOW-POWER SENSOR APPLICATIONS AT MICROWAVE BANDS. AUTOR:. Antonio Alex Amor. TUTOR:. José Manuel Fernández González. COTUTOR:. Pablo Padilla de la Torre. DEPARTAMENTO: Señales, Sistemas y Radiocomunicaciones. MIEMBROS DEL TRIBUNAL CALIFICADOR. PRESIDENTE: VOCAL: SECRETARIO: SUPLENTE:. FECHA DE LECTURA: CALIFICACIÓN:.

(4) Contents 1 Introduction 1.1 Energy Harvesting. Full System Architecture . . . . 1.2 Radio Spectrum. Bands of Interest . . . . . . . . . 1.3 Project Description . . . . . . . . . . . . . . . . . . 1.4 State of the Art of RF Energy Harvesting Systems .. . . . .. . . . .. . . . .. . . . .. . . . .. . . . .. . . . .. . . . .. . . . .. 2 Radiofrequency Harvester 2.1 Design of an Archimedean Spiral Antenna . . . . . . . . . . . . . . 2.2 Comparison between a Multiband PIFA and an Ultrawideband Archimedean Spiral Antenna . . . . . . . . . . . . . . . . . . . . . . 2.2.1 New Design . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3 Miniaturization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4 Microstrip to Parallel Strip Balun. Ultrawideband Design . . . . . . 2.4.1 Back-to-Back . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4.2 Balun-Antenna Connection . . . . . . . . . . . . . . . . . . 2.5 Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.5.1 Reflection Coefficient Measurement . . . . . . . . . . . . . . 2.5.2 Power Spectrum Measurement . . . . . . . . . . . . . . . . .. . . . .. 14 14 15 15 17. 20 . 20 . . . . . . . . .. 24 26 29 32 35 36 37 37 38. 3 Conditioning Circuit. Cockcroft-Walton Multiplier 42 3.1 Design of the Half-Wave Cockcroft-Walton Multiplier . . . . . . . . . 42 3.2 Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44 4 Matching Circuit 4.1 Narrow-band modeling of the Archimedean spiral antenna 4.2 Equivalent Circuit Model. Test Circuits . . . . . . . . . . . 4.2.1 First Model. Test Circuit #1 . . . . . . . . . . . . 4.2.2 Second Model. Test Circuit #2 . . . . . . . . . . . 4.2.3 Third Model. Test Circuit #3 . . . . . . . . . . . . 4.3 Nonlinearity Effects. Circuit #3 . . . . . . . . . . . . . . . 4.4 Comparison Among Different Number of Stages . . . . . .. . . . . . . .. . . . . . . .. . . . . . . .. . . . . . . .. . . . . . . .. . . . . . . .. 48 49 51 52 53 56 57 58. 5 Conclusion & Future Work 62 5.1 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62 5.2 Future Work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63. 4.

(5) CONTENTS. 5. A Harmonic Balance Method 65 A.1 A Simple Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66 B Design of L-Matching Networks 68 B.1 Matching Circuit in the Energy Harvesting System . . . . . . . . . . 69 C Reflection on Ethical, Economic, Social and Environmental Aspects 70 D Cost Estimation. 73. E Publications. 75.

(6) List of Figures 1.1 1.2 1.3. 1.4 1.5 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 2.10 2.11 2.12. 2.13 2.14. Full RF energy harvesting system. . . . . . . . . . . . . . . . . . . . Section of the electromagnetic spectrum. Source: [4]. . . . . . . . . Three-Element Dual-Band Yagi Rectenna (b) and a single element (equiangular spiral antenna) of the rectenna array implemented in [3] (b). Sources: [9] and [3], respectively. . . . . . . . . . . . . . . . . . Simulated range of optimal source impedances for one of the diodes used in [3]. Source: [3]. . . . . . . . . . . . . . . . . . . . . . . . . . Full-wave rectenna shown in [12](a), and a rectenna based on a Cockcroft-Walton multiplier [13](b). . . . . . . . . . . . . . . . . . .. . 15 . 16. . 17 . 18 . 19. Current distribution over a miniaturized Archimedean spiral antenna. Archimedean spiral antenna. . . . . . . . . . . . . . . . . . . . . . . Reflection coefficient of the Archimedean spiral antenna (normalized to 188.5 Ω). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Axial ratio (AR) of the Archimedean spiral antenna at different frequencies in a cut in the plane φ = 90o . . . . . . . . . . . . . . . . . . Total efficiency at different frequencies of the Archimedean spiral antenna. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Farfield radiation pattern at different frequencies of the Archimedean spiral antenna. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Parts of a dual-band PIFA antenna. . . . . . . . . . . . . . . . . . . . Reflection coefficient of the dual-band PIFA antenna. . . . . . . . . . Tri-band PIFA antenna. . . . . . . . . . . . . . . . . . . . . . . . . . Transversal cut of the proposed multiband PIFA (a) and its top (b) and bottom (c) planes. . . . . . . . . . . . . . . . . . . . . . . . . . . Inclination of PIFA’s patches. . . . . . . . . . . . . . . . . . . . . . . Monte Carlo simulation on the reflection coefficient of the proposed multiband antenna when varying ±2o the inclination of the plane of both PIFAs. The purple thick line represents the ideal scenario (null inclination) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Farfield radiation pattern at different frequencies of PIFA antenna. . . Different designs of the Archimedean spiral antenna: (a) no miniaturization [14], (b) miniaturization without impedance step, and (c) miniaturization with an impedance step. . . . . . . . . . . . . . . . .. 6. 21 22 22 23 23 24 25 26 26 27 27. 27 28. 30.

(7) LIST OF FIGURES 2.15 Simulated reflection coefficient of the Archimedean spiral antenna (normalized to 188.5 Ω) with different designs: (a) no miniaturization [14], (b) miniaturization without impedance step, and (c) miniaturization with an impedance step (20.72 x 19.77 cm vs 15.22 x 14.27 cm). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.16 Simulated efficiencies (radiation efficiency and total efficiency) of the miniaturized (one impedance step) Archimedean spiral antenna, and two examples of the spiral’s 3D radiation pattern at two different frequencies of interest. . . . . . . . . . . . . . . . . . . . . . . . . . 2.17 Simulated axial ratio, at different frequencies, of the Archimedean spiral antenna in two cuts in the planes φ = 0o (continuous line) and φ = 90o (dashed line). . . . . . . . . . . . . . . . . . . . . . . . . . 2.18 Simulated farfield patterns of the miniaturized Archimedean spiral antenna at the different frequencies of interest on the (a) vertical and (b) the horizontal plane. . . . . . . . . . . . . . . . . . . . . . . . . 2.19 Ultrawideband microstrip to parallel strip balun. . . . . . . . . . . 2.20 Electric field in the parallel strips’ port, and the QTEM (a) and the evanescent hybrid (b) modes. . . . . . . . . . . . . . . . . . . . . . 2.21 Simulated S-parameters of the microstrip to parallel strips balun. Note how the hybrid mode is quite attenuated with respect to the QTEM mode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.22 Back-to-back configuration of the UWB microstrip to parallel strip balun. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.23 Simulated S-parameters of the microstrip to parallel strips back-toback configuration. . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.24 Balun-antenna connection. . . . . . . . . . . . . . . . . . . . . . . . 2.25 Reflection coefficient of the joint structure (antenna + balun). . . . 2.26 Prototype of the miniaturized Archimedean spiral antenna. . . . . 2.27 Comparison of the simulated and measured reflection coefficients of the Archimedean spiral antenna (normalized to 188.5 Ω). . . . . . 2.28 Power spectrum measured by the Archimedean spiral antenna inside and outside the laboratory with the most relevant bands remarked: 1-FM, 2-DTT, 3-LTE-800, 4-GSM-900, 5-GSM-1800, 6-LTE-2100, 7WiFi, 8-LTE-2600, 9-WiFi. . . . . . . . . . . . . . . . . . . . . . . 2.29 Measured reflection coefficient of the manufactured 2.4-GHz patches (Dimensions: small patch: 4.89 x 3.91 cm, circularly-polarized patch: 5.05 x 3.91 cm, big patch: 5.02 x 4.07 cm. . . . . . . . . . . . . . . 2.30 Comparison between the power spectrum measured by the Archimedean spiral antenna inside the laboratory and by the three 2.4-GHz patches, with the most relevant bands remarked: 1-FM, 2DTT, 3-LTE-800, 4-GSM-900, 5-GSM-1800, 6-LTE-2100, 7-WiFi, 8LTE-2600, 9-WiFi. . . . . . . . . . . . . . . . . . . . . . . . . . . .. 7. . 30. . 31. . 31. . 32 . 33 . 34. . 34 . 35 . . . .. 36 36 37 37. . 38. . 39. . 40. . 40.

(8) LIST OF FIGURES 3.1. 3.2. 3.3 3.4 3.5 3.6 4.1. 4.2. 4.3. 4.4 4.5. 4.6. Circuital scheme of the n-stage half-wave Cockcroft-Walton voltage multiplier: the Villard circuit and the half-wave rectifier are circled in orange and in red, respectively, and the Greinacher voltage doubler is remarked in dashed blue line. . . . . . . . . . . . . . . . . . . . . . HWCW test set ((a) upper side, (b) bottom side) with six different combinations of components and number of stages: 33 nF – HSMS 2822 – 3 stages (blue), 33 nF// 33 pF – HSMS 2822 – 2 stages (purple), 33 pF – HSMS 2850 – 2 stages (green), 33 pF – HSMS 2822 – 2 stages (orange), 33 pF – HSMS 2850 (red) – 2 stages, 33 pF – HSMS 2822 – 5 stages (cyan). . . . . . . . . . . . . . . . . . . . . . . . . . . Measurement set-up: Agilent E4438C vector signal generator, Yokogawa DL9240L digital oscilloscope, HWCW test set, and a multimeter. Output DC voltages in a 2-stage HWCW scheme when using different components (no-load situation). . . . . . . . . . . . . . . . . . . . . . Output voltage versus frequency in the 2-stage HWCW scheme (with C=33 pF) at different input powers. . . . . . . . . . . . . . . . . . . . Output voltage at each state of the 5-stage HWCW scheme (with C=33 pF) for an input power of -10 dBm. . . . . . . . . . . . . . . . General scheme of the RF harvesting system. Note as the source impedance Zs depends on five terms: the input power Pin , the input range of frequencies f , the number of stages N in the CockcroftWalton multiplier , the intrinsic impendance of the antenna Zant , and the load ZL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Scheme of the RF harvesting system particularized to the use of the Archimedean spiral antenna. Notes: *Pin1 = −5.80 dBm and Pin1 = −1.84 dBm, inside and outside the laboratory, respectively; and Pin2 = −9.20 dBm and Pin1 = −2.34 dBm, inside and outside the laboratory, respectively. **fin1 = 807 MHz, and fin2 = 942 MHz. Zoom over the power spectrum measured by the Archimedean spiral antenna at 800 and 900 MHz. The delta functions marked in blue and orange represent the modeling of the Archimedean spiral antenna as a generator, inside and outside the laboratory, respectively. . . . . Narrow-band circuital model of the antenna. . . . . . . . . . . . . . Test circuit #1 (a) and its equivalent model (b). The source impedance that maximizes the power delivered to the load is Zs = 60 + j200 Ω. Values of the components: Lmat = 33 nH, C = 33 pF, RL = 6.1 kΩ. Values of the parasitics: Cp = 0.77 pF, Cpd = 0.70 pF, Lp = 2.64 nH. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Measurements relative to test circuit #1 (blue line) and the simulations of the circuit without parasitic elements (red line) and the first circuit model (magenta line). Cp = 0.77 pF, Cpd = 0.70 pF, Lp = 2.64 nH. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 8. 42. 44 45 46 47 47. 48. 49. 50 50. 52. 53.

(9) LIST OF FIGURES 4.7. 4.8. 4.9 4.10. 4.11. 4.12 4.13 4.14 4.15. Test circuit #2 (a), first model applied on test circuit #2 (b), and the final equivalent model (c) that takes into account the parasitic elements of the board. The source impedance that maximizes the power delivered to the load is Zs = 50 + j78 Ω. Values of the components: Lmat1 = 4.7 nH, Lmat2 = 8.2 nH, C = 33 pF, RL = 2.34 kΩ. Values of the parasitics: Cp1 = 0.70 pF, Cp2 = 0.40 pF, Cpd1 = 0.70 pF, Cpd2 = 0.10 pF, Lpd = 1.5 nH, Lp = 1.5 nH, Lvia = 0.5 nH. . . . . . . Measurements relative to test circuit #2 (blue line) and the simulations of the circuit using the first circuit model (red line) and a redesigned second circuit model (gold line). . . . . . . . . . . . . . . . Measurement set-up of test circuit #2. . . . . . . . . . . . . . . . . . Circuit #3 (a) and the third model applied on it (b). The source impedance that maximizes the power delivered to the load is Zs = 18 + j56 Ω. Values of the components: Lmat1 = Lmat22 = 6.8 nH, Lmat21 = 3.3 nH, Cmat1 = 1.2 pF, Cmat21 = Cmat22 = 1.5 pF, C = 33 pF, RL = 2.34 kΩ. Values of the parasitics: Cp1 = Cp22 = 2.2 pF, Cp21 = 1.3 pF, Cpd1 = 0.75 pF, Cpd2 = 0.10 pF, Lpd = 1.5 nH, Lpd1 = Lpd2 = 0.5 nH, Lp = 1.5 nH, Lvia = 0.55 nH. . . . . . . . . . . Measurements relative to test circuit #3 (blue line) and the simulations of the circuit using the first circuit model (red line), the second circuit model (gold line), and a redesigned third model (purple line). . Nonlinearity in circuit #3 caused by different input power levels. . . . Measured efficiency in circuit #3 as a function of the input power. . . Efficiency of the simulated circuit that includes the optimum source impedances as a function of the load RL . . . . . . . . . . . . . . . . . Efficiency of the simulated circuit that implements the final Lmatching networks as a function of the load RL . . . . . . . . . . . . .. 9. 54. 54 55. 56. 57 58 58 60 61. A.1 Partitioning of a microwave circuit into linear and non-linear subcircuits. The source Zs (ω) and load ZL (ω) impedances have been placed into the linear circuit. The circumflex accent indicates a non-linear term. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65 A.2 A diode D connected to a lumped element with an sinusoidal excitation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66 B.1 The two possible configurations of an L-matching network. . . . . . . 69.

(10) List of Tables 2.1. Measurements on the harvested power. . . . . . . . . . . . . . . . . . 41. 4.1. Simulated optimum source impedances Zs , load resistances RL , output voltages Vo , output currents Io , output powers Po , and efficiencies for the different number of stages of the multiplier circuit with the ”outside” modeling of the antenna. . . . . . . . . . . . . . . . . . . . 59 Simulated optimum source impedances Zs , load resistances RL , output voltages Vo , output currents Io , output powers Po , and efficiencies for the different number of stages of the multiplier circuit with the ”inside” modeling of the antenna. . . . . . . . . . . . . . . . . . . . . 59. 4.2. D.1 Cost Estimation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74. 10.

(11) List of Abbreviations and Acronyms.. AC ADS AR CMOS DC DCR DTT ECG EM ESR EU FM FR4 FSA GSM HB HWCW IoT LTE PIFA PCB RF SMA SMT SRF UWB VNA WiMAX WSN. — — — — — — — — — — — — — — — — — — — — — — — — — — — — —. Alternating Current Advanced Design System Axial Ratio Complementary Metal-Oxide-Semiconductor Direct Current Direct Current Resistance Digital Terrestrial Television Electrocardiogram Electromagnetic Equivalent Series Resistance European Union Frequency Modulation Flame-Retardant 4 Fibonacci Spiral Antenna Global System for Mobile Communications Harmonic Balance Half-Wave Cockcroft-Walton Internet of Things Long Term Evolution Planar Inverted-F Antenna Printed Circuit Board Radiofrequency SubMiniature version A Surface-Mount Technology Self-Resonant Frequency Ultrawideband Vector Network Analyzer Worldwide Interoperability for Microwave Access Wireless Sensor Network. 11.

(12) Design of a Radiofrequency (RF) Energy Harvesting System for Low-Power Sensor Applications at Microwave Bands Antonio Alex Amor Keywords: energy harvesting, antenna design, Archimedean spiral antenna, ultrawideband, matching circuit, Cockcroft-Walton multiplier, storage circuit.. Abstract Energy harvesting is defined as the process of obtaining energy from external sources, such as wind, sunlight, heat or radiofrequency (RF) transmitters; which is later conveniently processed and stored in order to feed, in general, low-power electronic systems. Generally, a RF energy harvesting system is formed by four stages: • RF harvester: an antenna capable of acquiring RF ambient power from the bands of interest of the radio spectrum. • Conditioning circuit: a circuit that converts the signal acquired by the antenna, which is a sum of radio spectrum carriers, into a DC supply. • Matching circuit: it is situated between RF harvester and conditioning circuit stages. It maximizes the power delivered from the antenna to the conditioning circuit. • Storage circuit: it stores the available energy in order to feed an electronic system (typically a sensor). The aim of this project is the design of a radiofrequency energy harvesting system as a “free” power source for low-power electronic systems. The work is presented with the following phases and main development tasks: Firstly, a set of different antennas are being studied. We will focus this chapter on the study of multiband, wideband, ultrawideband, and frequency independent antennas. Simulations will be carried out with CST Microwave Studio Suite, in order to optimize the design that will later be manufactured and measured. Secondly, we will focus on the conditioning circuit. One of the topologies of our interest is the Cockcroft-Walton voltage multiplier, a circuit that rectifies and elevates the signal acquired by the antenna. We will simulate and test different element configurations on it. Thirdly, we will maximize the power transfer (from the antenna to the circuit) by placing an impedance transformer circuit. We must study which is the.

(13) LIST OF TABLES. 13. source impedance (the receiver antenna can be seen as a generator with a source impedance) that maximizes the output power, and then transform the antenna impedance into this one. For this task, we will lay on the use of simulators that implement “harmonic balance”, a technique commonly used to calculate the steady-state response of nonlinear differential equations, and it is mostly applied to nonlinear circuits (as the Cockcroft-Walton multiplier). Finally, we will design an energy storage circuit in order to conveniently store and manage the previously conditioned power. It actually serves as a battery for the low-power electronic system to be fed. Supercapacitors (capacitors of units or tens of Farads) seem as the best candidate for this purpose..

(14) Chapter 1 Introduction 1.1. Energy Harvesting. Full System Architecture. Energy harvesting is defined as the process of obtaining energy from external sources, such as wind, sunlight, heat or radiofrequency (RF) waves; which is later conveniently processed and stored in order to feed, in general, low-power electronic systems. The most common energy harvesting techniques [1] are based on the reuse of the sunlight; on the reuse of kinetic energy, through mechanic movements or deformations inside the device; on the reuse of thermal energy, through thermocouples that take advantage of the Seebeck effect [1]; and on the capture of electromagnetic radiation. The constant increase in the number of RF transmitters has led to a non-negligible level of power at some particular frequency bands. This fact, joined to the reduction in the power consumption necessities of many devices, has turned RF energy harvesting into a feasible energy source for such low-consumption devices. For instance, as discussed in [2], several harvesting devices are starting to be used for medical purposes, in order to measure intraocular pressure, temperature, electrocardiograms (ECG), etc.; or in fields as wireless sensor networks (WSN) and Internet of Things (IoT). Most of these devices consume less than 30 µW. Typically, a RF energy harvesting system is formed by four stages [3], as shown in Fig. 1.1. • RF harvester: an antenna capable of acquiring RF ambient power from the bands of interest within the radio spectrum. • Conditioning circuit: a circuit that converts the signal acquired by the antenna, which is a sum of radio spectrum carriers, into a DC supply. • Matching circuit: it is situated between RF harvester and conditioning circuit stages. It maximizes the power delivered to the conditioning circuit. • Storage circuit: it stores the acquired power in order to feed a certain electronic system (typically a sensor). 14.

(15) CHAPTER 1. INTRODUCTION. 15. Fig. 1.1: Full RF energy harvesting system.. 1.2. Radio Spectrum. Bands of Interest. We name by electromagnetic spectrum to the range of all possible electromagnetic radiations. It extends approximately from radio long waves to gamma rays. The wavelength λ of a disturbance and the frequency of it, f , are linked by the inversely proportional relationship c (1.1) λ= f where c is the propagation speed of the wave in a certain medium. Fig. 1.2 presents a section of the electromagnetic spectrum as a function of the frequency/wavelength (top/bottom), and the most typical applications withing the radio spectrum. In the vast majority of countries, almost all relevant RF commercial applications are located between 80 MHz and 5 GHz. In our particular case, we should pay special attention to 500, 800, 900, 1800, 2400, 3500 and 5000 MHz bands [5–7], where DTT (Digital Terrestrial Television), LTE (Long Term Evolution), GSM (Global System for Mobile Communications), WiFi and WiMAX (Worldwide Interoperability for Microwave Access) are located. Another important power contribution comes from FM band (80 - 100 MHz), but note the size of the antenna becomes quite elevated to be considered here.. 1.3. Project Description. This project presents a radiofrequency ultrawideband energy harvesting system, based on an ultrawideband planar antenna that captures the spectrum energy, connected to a voltage multiplier circuit that rectifies the signal and converts it into DC power supply. The document is organized as follows:.

(16) CHAPTER 1. INTRODUCTION. 16. Fig. 1.2: Section of the electromagnetic spectrum. Source: [4]. • Chapter 1 introduces the global design of the proposed RF energy harvesting system and its working scheme. • Chapter 2 presents the design of the radiofrequency harvester. Firstly, a comparison between multiband and wideband antennas is made, coming to the conclusion that it is usually wiser using wideband antennas instead of multiband ones. Secondly, we focus on the study, design and simulation of the Archimedean spiral antenna and the ultrawideband balun required to the measurement of the antenna. Thirdly, several miniaturization techniques are investigated to try reducing the physical dimensions of the spiral. • Chapter 3 depicts the design of the conditioning circuit, based on a half-wave Cockcroft-Walton multiplier. Besides that, a comparison between different low forward voltage Schottky diodes and RF capacitors is made in order to reduce voltage ripple and losses of the DC output signal. • Chapter 4 studies the matching circuit. It is analyzed the effect of modifying both the input power (power acquired by the antenna) and the load (sensor).

(17) CHAPTER 1. INTRODUCTION. 17. connected to the conditioning circuit in order to maximize the power delivered from the antenna to the Cockcroft-Walton multiplier circuit. • Chapter 5 summarizes the work and points out the main conclusions extracted from it.. 1.4. State of the Art of RF Energy Harvesting Systems. In this section, we illustrate the typical steps carried out in the design of each of the four stages that form the RF harvesting system, and the most typical problems derived from them. We also review the current RF energy harvesting systems, showing the most typically used antennas, frequency bands, and conditioning circuits. The two most commonly used types of antennas as harvesting elements are multiband and wideband antennas. Multiband antennas are usually narrow-band, and they are often easier to design due to their well-known performance. To achieve multiple resonances, it is frequent to make cuts or slots in the structure, which can be a double-edged sword since can be difficult to control the mutual coupling between the slots themselves. On the other hand, wideband antennas can be harder to design, although it depends on the situation. In return, its broadband behavior assures as that the antenna certainly covers the entire region of interest. Fig. 1.3 shows two different rectenna (rectifying antennas) prototypes: one of them is based on the three-element dual-band Yagi antenna presented in [9] (Fig. 1.3a), which operates at 945 MHz and 2.45 GHz; and the other one is based on a equiangular spiral antenna developed in [3] that covers from 2 GHz to 18 GHz (Fig. 1.3b).. (a) Three-element dual-band Yagi rectenna. (b) Equiangular spiral. Fig. 1.3: Three-Element Dual-Band Yagi Rectenna (b) and a single element (equiangular spiral antenna) of the rectenna array implemented in [3] (b). Sources: [9] and [3], respectively..

(18) CHAPTER 1. INTRODUCTION. 18. The most common, but smarter, circuit approximation to the design of the system consists in finding the optimum source impedance that maximizes the efficiency of the conditioning circuit. For example, 1.4 shows the simulated range of normalized optimum source impedances over the Smith chart for a variety of studied cases: high input power and low frequency (A), low input power and low frequency (B), low input power and high frequency (C), high input power and high frequency (D). Note that the shaded area is large, which indicates that the circuit has a strongly non-linear behavior, that is, its performance depends on the input power and other factors. Nevertheless, one problem is that the impedance of the antenna does not usually coincide with the optimum source impedance in the operating frequency range, so a matching network is typically needed between the antenna and the conditioning circuit, as Fig. 1.1 illustrates. The current bottleneck of RF harvesting systems is the conditioning circuit, which is in charge of eliminating the AC level and rectifying the input signal. This is due to the intrinsic consumption of the diodes placed in the circuit (they are not ideal), which at low input power levels dissipate a high relative part of the total power. In consequence, for input powers of less than 1 mW, the efficiency of a typical half/fullwave rectifier, defined as the proportion between the output DC power and the input RF power η(%) =. PoutDC · 100, PinRF. (1.2). Fig. 1.4: Simulated range of optimal source impedances for one of the diodes used in [3]. Source: [3]..

(19) CHAPTER 1. INTRODUCTION. 19. is typically under 30%. In [3], a half-wave rectifier circuit is implemented over an array of equiangular spirals. The results show that the efficiency of the rectifier ranges from 1% to 20%, depending on the input power level. Some recent (and very advanced) studies such as [11] have achieved efficiencies of up to 74% for 1 mW input power, by externally feeding the circuit in question (CMOS configuration) and using the self-body biasing technique, which allows us to change the threshold voltage Vt and turn on quicker the transistor. Nevertheless, all schemes that reach high efficiencies in the rectifier circuit are not ”passive”, that is, they are externally powered. The most used configuration for a conditioning circuit is the half-wave rectifier, but there are other more complex configurations that are worth mentioning. For example, [12] presents a full-wave rectenna, formed by two concentric squared patches attached to a full-wave rectifier (Fig. 1.5a). The results show an efficiency in the rectifier circuit of less than 34% for an input power of 1 mW. Another passive (not externally biased) configurations arise from multiplier circuits. They are capable of rectifying the signal and elevating the output voltage at the same time, which is of great interest in our application since the input voltages are of the order of hundreds of mV. In these terms, [13] presents a rectenna based on a coplanar broadband antenna and a single stage of a Cockcroft-Walton multiplier.. (a) Full-wave rectenna. Source: [12]. (b) Rectenna based on a Cockcroft-Walton multiplier. Source [13]. Fig. 1.5: Full-wave rectenna shown in [12](a), and a rectenna based on a CockcroftWalton multiplier [13](b)..

(20) Chapter 2 Radiofrequency Harvester When choosing which type of spiral to use, factors such as the purity of the circular polarization or the bandwidth of the spiral should be taken into account. In [16], a comparison between equiangular and Archimedean spiral antennas is made. It is pointed out that the Archimedean spiral shows an improved axial ratio and a wider bandwidth for a given outer diameter. The design and manufacturing of the Archimedes spiral is also easier than in the case of the equiangular spiral, due to the constant angular increase of its arms. Besides that, wideband spiral antennas may be configured with single, double or multiple arms [17], the multiple-arm spiral antenna being a good option to provide squinted beams. In the light of previous works [16,17], the two-arms Archimedean spiral antenna seems as the proper option to be implemented in practice.. 2.1. Design of an Archimedean Spiral Antenna. The Archimedean spiral is typically classified within the group of frequency independent antennas, that is, antennas that maintain some of their radiation parameters constant, as the input impedance or the bandwidth, in relation to the frequency. Victor H. Rumsey laid the foundations of this subject. According to him [15], those antennas that are entirely defined by angles, as the Archimedean spiral or the equiangular spiral, show a frequency independent behavior. Theoretically, an infinite-sized Archimedean spiral antenna in a self-complementary design (the arm width w is equal to the gap width s) presents a constant impedance that is fixed (according to Babinet’s principle) to the value Zant = Zcomp = 60π Ω ≈ 188.5 Ω,. (2.1). where Zcomp is the impedance of its complement. The finite size of an Archimedes antenna causes a finite bandwidth, which becomes more reduced as the antenna becomes smaller, with its impedance varying nearby 188.5 Ω. This fact can be noticed in Fig. 2.27. On the other hand, several studies emphasize [3, 14, 18, 19] that the current distribution over the surface of the Archimedean spiral is concentrated close to the center 20.

(21) CHAPTER 2. RADIOFREQUENCY HARVESTER. (a) 10 GHz. (b) 500 MHz. 21. (c) 300 MHz. Fig. 2.1: Current distribution over a miniaturized Archimedean spiral antenna. of the antenna at high frequencies, and opens to its end at low frequencies. This current distribution (Fig. 2.1) leads to two equations that estimate the dimensions of the antenna according to the desired bandwidth (determined by the lower and the upper cutoff frequencies, fL and fH , respectively). These equations are presented as follows co , (2.2) r1 = √ 2πfH εref f co r2 = , (2.3) √ 2πfL εref f where r1 is the inner radius, r2 is the outer radius, co is the speed of light in the vacuum, and εref f is the effective relative permittivity of the dielectric used (FR4: εr = 4.7, tan δ = 0.014 @ 1MHz). Due to the complex geometry of the antenna, this dimension estimation is revealed as a good starting point to model the Archimedes spiral on a full-wave electromagnetic simulator. As no ground plane is used, it is really difficult to estimate in an accurate manner εref f , so it is taken εr2+1 = 2.85 as the best approximation to it. Self-complementarity is only reachable if the metallic strip width, w, is equal to the separation between two adjacent strips, s, that is, w = s. Fixing the upper and lower cutoff frequencies as 18 GHz and 0.4 GHz, respectively, and applying (2.2) and (2.3), it was found that the inner radius must be r1 = 1.5 mm and the outer radius r2 = 70.7 mm. Note that these values are mainly indicative, since, as seen in Fig. 2.1, not all the current distribution over the surface of the spiral is located at the center or at the end of the antenna. In fact, as will be seen later, equation (2.3) works in a worse manner than equation (2.2), due to the current distribution at low frequencies does not really move to the end of the antenna, but to an intermediate strip. Therefore, the real value of r2 will be higher than the one calculated in (2.3). In Fig. 2.2, it is presented the Archimedean spiral antenna, designed over single-sided FR4 copper clad (no ground plane). As can be noticed, the value of the inner radius is r1 = 1.20 mm, similar to the calculated value in equation (2.2),.

(22) CHAPTER 2. RADIOFREQUENCY HARVESTER. 22. Fig. 2.2: Archimedean spiral antenna. but the outer radius is slightly higher, with a value of r2 = 97.5 mm. On the other hand, the metallic strip width w and the gap s were estimated by numerical simulations, giving us a value of w = s = 3.9 mm. Figure 2.3 shows the reflection coefficient of the Archimedean spiral antenna (normalized to 188.5 Ω). The antenna offers a bandwidth of 17.57 GHz (|S11 | < −10 dB), which ranges from 0.44 GHz to 18.01 GHz. It covers DTT, LTE, GSM, WiFi and WiMAX bands, and it is circularly polarized (AR < 3 dB) in all bands but DTT, where this value is surpassed, as depicted in Fig. 2.4. A ripple in the lower frequencies of the reflection coefficient is also observed, product of the short electrical size of the antenna. This means that the Archimedean spiral begins to show a resonant behavior and its impedance value is no longer constant at the lower frequencies.. Fig. 2.3: Reflection coefficient of the Archimedean spiral antenna (normalized to 188.5 Ω)..

(23) CHAPTER 2. RADIOFREQUENCY HARVESTER. 23. Fig. 2.4: Axial ratio (AR) of the Archimedean spiral antenna at different frequencies in a cut in the plane φ = 90o . In Fig. 2.5, it is plotted the total efficiency of the Archimedean spiral antenna. As can be seen, the efficiency of the spiral is superior to 82% (0.82 if we normalize it to the unity) in all the bands of interest. Furthermore, the farfield radiation pattern displayed in Fig. 2.6 shows the omnidirectionality of the antenna and its backward radiation (due to no ground plane is used). Omnidirectionality and circular polarization are key factors in RF energy harvesting, due to we do not know the polarization and the direction of arrival of the incoming energy. Besides that, note that the directivity of the antenna increases slightly with frequency, having 3.57 dBi at 830 MHz and 6.15 dBi at 2600 MHz.. Fig. 2.5: Total efficiency at different frequencies of the Archimedean spiral antenna..

(24) CHAPTER 2. RADIOFREQUENCY HARVESTER. 24. Fig. 2.6: Farfield radiation pattern at different frequencies of the Archimedean spiral antenna.. 2.2. Comparison between a Multiband PIFA and an Ultrawideband Archimedean Spiral Antenna. The PIFA (Planar Inverted F-Antenna) is a type of printed antenna commonly used in mobile communications. It can be considered as a particularization of a half-wave patch, where its center (the electric field is null at the center of a λ/2 patch) is short-circuited to the ground plane. This technique is very smart and allows reducing the size of the PIFA to more than half (≤ λ/4) without degrading any of its parameters, which permits placing it in small spaces, as mobile phones. The antenna presents a low directivity (3 – 8 dBi) and a high efficiency (around 80 %), so, combined with some appropriate slots over its surface, it can be transformed into a multiband antenna. Nevertheless, it is linearly polarized and as we do not know the polarization of the incoming signals, the depolarization losses will be higher than in the case of a circularly-polarized antenna. The design equation of the PIFA antenna is presented in [21] and has the form.

(25) CHAPTER 2. RADIOFREQUENCY HARVESTER. 25. λd , (2.4) 4 where L is the length, W is the width, Ws the short-circuited stub width, and λd the operating wavelength of the PIFA antenna. Note that reducing Ws permits us reducing the size of the antenna (L, W ), but the lower the stub width, the lower the antenna bandwidth (8 %). If we design the PIFA to resonate at 930 MHz (without any dielectric, in order to reduce losses), and we assume a stub width of Ws = 7 mm and a relation width-length W/L = 0.95, the length and the width of the PIFA must be L = 44.9 mm and W = 42.7 mm, respectively. L + W − Ws =. In Fig. 2.7, it is presented a typical dual-band GSM PIFA. It is formed by the elements described above (quarter-wave patch, ground plane, short-circuited stub), which permit the antenna to resonate at 930 MHz, and a rectangular cut that makes the antenna to resonate at 1830 MHz. No dielectric is used in order to reduce at maximum the losses. The placement of the feeding probe as well as the aperture of the ground plane are key factors in order to match the antenna. Note that when the coaxial probe moves towards the short-circuited stub, the input impedance of the antenna goes to zero (the impedance of a short-circuited stub is zero). In Fig. 2.8, it is shown the reflection coefficient of the dual-band GSM PIFA. It covers the 930 and the 1830 MHz bands with a narrow band. Note as well that the matching at 930 MHz can be improved, although the -6 dB requirement in mobile communications is fulfilled.. Fig. 2.7: Parts of a dual-band PIFA antenna..

(26) CHAPTER 2. RADIOFREQUENCY HARVESTER. 26. Fig. 2.8: Reflection coefficient of the dual-band PIFA antenna.. 2.2.1. New Design. More resonances can be achieved by adding U slots, as shown in Fig. 2.9. But the fact is that adding several slots greatly hinders the design of PIFA due to the mutual coupling between elements. Thus, it is not recommended to try to place more than three slots in the same patch.. Fig. 2.9: Tri-band PIFA antenna. According to the commented before, we propose a new design where two different PIFAs are integrated in the same ground plane (90 x 55 mm), which is fed with a single coaxial probe (SMA connector) through a distribution network printed in a FR4 substrate (εr = 4.7, tan δ = 0.014 @ 1MHz) and situated below the ground plane, as shown in Fig. 2.10. The feeding of the new design is more complicated than in the previous case, but in return the antenna can reach more bands. Two metallic vias cross the structure, from the end of the distribution network to the feed points in both PIFAs, in order to feed the antenna. As mentioned in Fig. 2.10, the PIFA placed on the right side of the image covers all GSM bands, while the other covers LTE-2100, WiFi, and LTE-2600 bands..

(27) CHAPTER 2. RADIOFREQUENCY HARVESTER. (a) Transversal Cut. 27. (b) Top Plane. (c) Bottom Plane. Fig. 2.10: Transversal cut of the proposed multiband PIFA (a) and its top (b) and bottom (c) planes.. Fig. 2.11: Inclination of PIFA’s patches.. Fig. 2.12: Monte Carlo simulation on the reflection coefficient of the proposed multiband antenna when varying ±2o the inclination of the plane of both PIFAs. The purple thick line represents the ideal scenario (null inclination).

(28) CHAPTER 2. RADIOFREQUENCY HARVESTER. 28. In Fig. 2.12, it is presented a Monte Carlo simulation of the reflection coefficient in the proposed multiband antenna, where the inclination of the plane of both PIFAs has set to move in a range of ±2o (look at Fig. 2.11). The purple thick line represents the ideal scenario (null inclination). Note that there are four resonances in Fig. 2.12: the first of them caused by PIFA #1; the second one (1450 MHz) caused by a parasitic resonance either of the ground plane or the metallic vias or the distribution network; the third one (1830 MHz) caused by an appropriate cut in PIFA #1; and the fourth one (2100-2700 MHz) caused by a resonance of PIFA #2. As Fig. 2.12 proves, a slight manufacturing deviation on the inclination of both PIFAs (≤ 2o ) may mismatch the element in the bands of interest, leaving the antenna inoperative. Fig. 2.13 shows the far field radiation pattern of the proposed antenna at different frequencies. As expected, the multiband PIFA presents a non-directive behavior. Its total efficiency varies along the frequency according to its radiation efficiency and the antenna matching, being on average in a 76 % (the maximum is ubicated at 930 MHz, with a 95 %, and the minimum at 1830 MHz, with a 55 %). In summarize, we have designed and simulated a multiband PIFA and we have corroborated that is preferable working with wideband antennas: they are less sensible to slight manufacturing deviations and they are interoperable among countries, where frequency assignment plans can vary.. Fig. 2.13: Farfield radiation pattern at different frequencies of PIFA antenna..

(29) CHAPTER 2. RADIOFREQUENCY HARVESTER. 2.3. 29. Miniaturization. In order to increase the RF harvested power per unit area (µW/cm2 ), it is essential to try to reduce the physical dimensions of the antenna. Several miniaturization techniques have been studied so far, most of them based on structural modifications and lumped element loadings [20, 24, 25]. Recent studies have tried to reduce the size of spiral antennas by using fractal geometries, as in [26], where a modified Koch curve is used to shrink a Fibonacci spiral antenna (FSA). We propose increasing the bandwidth of the Archimedean spiral by extending the arms to the end of the antenna (maintaining the same PCB area). Consequently, the electrical size of the Archimedes antenna is higher and therefore, it reaches lower frequencies without significant performance degradation. Fig. 2.14 depicts different designs following that approach. As the simulation results probe (Fig. 2.15), it is preferable to place an impedance step (Fig. 2.14c) when extending the arms of the antenna, so that reflections are softened and more bandwidth is achieved (fL1−step = 350 MHz vs fL0−step = 400 MHz vs fLnomin = 510 MHz). In these terms, Fig. 2 (purple line) shows that the area of the miniaturized antenna can be reduced 5.5 x 5.5 cm2 to meet the lower cutoff frequency (510 MHz) of the non-miniaturized antenna, that is, the area could be reduced a 7.38 % without appreciable performance degradation of the antenna. In Fig. 2.16, the radiation efficiency and the total efficiency of the miniaturized Archimedes antenna are plotted. The antenna shows a good efficiency behavior, with values over the 86% at all the frequencies of interest. It also was included in Fig. 2.16 a comparison of the farfield radiation pattern at two different frequencies of interest: 830 MHz (LTE) and 2450 MHz (WiFi). As mentioned before, notice that the 3D radiation pattern tends to be slightly more directive as the frequency increases. Besides that, Fig. 2.17 presents the axial ratio of the antenna, at the different frequencies of interest, in two cuts in the planes φ = 0o and φ = 90o . As it can be seen, the spiral is circularly polarized in a wide range of elevation angles (-65o to 65o ) at most frequencies, but 500 MHz, where it could not go under 3 dB and remains elliptically polarized..

(30) CHAPTER 2. RADIOFREQUENCY HARVESTER. (a) No miniaturization. 30. (b) 0-step minituarization. (c) 1-step miniaturization. Fig. 2.14: Different designs of the Archimedean spiral antenna: (a) no miniaturization [14], (b) miniaturization without impedance step, and (c) miniaturization with an impedance step.. Fig. 2.15: Simulated reflection coefficient of the Archimedean spiral antenna (normalized to 188.5 Ω) with different designs: (a) no miniaturization [14], (b) miniaturization without impedance step, and (c) miniaturization with an impedance step (20.72 x 19.77 cm vs 15.22 x 14.27 cm)..

(31) CHAPTER 2. RADIOFREQUENCY HARVESTER. 31. Fig. 2.16: Simulated efficiencies (radiation efficiency and total efficiency) of the miniaturized (one impedance step) Archimedean spiral antenna, and two examples of the spiral’s 3D radiation pattern at two different frequencies of interest.. Fig. 2.17: Simulated axial ratio, at different frequencies, of the Archimedean spiral antenna in two cuts in the planes φ = 0o (continuous line) and φ = 90o (dashed line). The farfield radiation pattern, in two cuts in the vertical and horizontal planes, is shown in Fig. 2.18. As depicted before, the antenna presents its maximum of radiation on the broadside direction, and since there is no ground plane, it also radiates symmetrically backwards. As the frequency increases, the 90o radiation null becomes more accentuated (Fig. 2.18a) and therefore the directivity of the antenna increases..

(32) CHAPTER 2. RADIOFREQUENCY HARVESTER. (a) Vertical Plane. 32. (b) Horizontal Plane. Fig. 2.18: Simulated farfield patterns of the miniaturized Archimedean spiral antenna at the different frequencies of interest on the (a) vertical and (b) the horizontal plane.. 2.4. Microstrip to Parallel Strip Balun. Ultrawideband Design. The two arms of the Archimedean spiral turn the antenna into a balanced structure and make feeding slightly difficult. This is not a problem in the case of placing the plus and the minus poles of the rectifier circuit directly attached to each arm of the antenna, because the balanced structure is preserved, but when the performance of the antenna is measured independently via an SMA (Subminiature version A) connector, the situation is quite different. The unbalance in the connection causes the currents in both arms to not be the same, that is, to not be symmetric. Subsequently, the radiation pattern changes and the antenna is no longer omnidirectional. Besides that, the input impedance of the Archimedean spiral is not referred to 50 Ω, but 188 Ω, as depicted in equation (2.1). A balun is an electrical device capable of transforming a signal from an unbalanced structure (such as a coaxial cable) to a balanced structure (such as a copper pair). There are many widely studied configurations [18], but most of them are narrow band solutions. However, different authors have implemented [27, 28] an ultrawideband microstrip to parallel strip balun oriented to use in spiral antennas, so that the impedance transformation (from 50 Ω to 188 Ω) is already contemplated. It is based on a tapered structure, which greatly increases the bandwidth. A typical design criterion is to set the length of the structure as λo /4 at the lowest operation frequency, thus, relative bandwidths of 200% are easily achievable. It is also interesting to use high-permittivity substrates in order to reduce the size of the balun, which can be significant whether we set the lower cutoff frequency at 300 MHz. So, we are working with Rogers 3006 substrate (εr = 6.15, tan δ = 0.0020 @10 GHz). In short, the challenge is threefold as commented above: transforming an unbalanced structure into a balanced one, and transforming the antenna impedance (188.

(33) CHAPTER 2. RADIOFREQUENCY HARVESTER. 33. Ω) into 50 Ω (SMA connector) within a ultra-wide range of frequencies (0.3-18 GHz). The design of the ultrawideban microstrip to parallel strip balun is shown in Fig. 2.19. The balun is formed by two metallic tapered strips placed in the top and bottom planes of a Rogers 3006 substrate. The wider section of the bottom strip acts as a ground plane for the top strip, so that the typical quasi-TEM mode of a microstrip line is propagated in the structure. The design equation for the metallic top strip is wt (x) = wtini eat x , (2.5) and for the bottom strip is wb (x) = wbini eab x ,. (2.6) 1 L. . wp wtini. . is the decay factor of the top strip, wbini is the initial width of the bottom strip, and ab = L1 ln wwb p ini is the decay factor of the bottom strip.. where wtini is the initial width of the top strip, at =. ln. Different simulation and optimization processes set the values mentioned before in: L = 200 mm, wtini = 1.8 mm, wbini = 12 mm, and wp = 0.15 mm. The initial width of the top strip wtini = 1.8 mm was calculated from the well-known microstrip equations [4, 18] in order to obtain 50 Ω in the input port for the chosen dielectric, while the initial width of the bottom strip wbini = 12 mm was obtained from a parametric sweep in CST Microwave Studio. The length of the balun L = 200 mm is about a quarter of wavelength at the lower operating frequency, 300 MHz. In Fig. 2.20, we can see the two modes present in the port relative to the parallel. Fig. 2.19: Ultrawideband microstrip to parallel strip balun..

(34) CHAPTER 2. RADIOFREQUENCY HARVESTER. 34. lines. The first one has a quasi-TEM structure, with the field lines emanating from the bottom strip and ending up on the top strip, ensuring a 180 degree phase shift within the two strips. On the other hand, the second mode is a non-desired evanescent hybrid mode that must suppressed. On the other hand, Fig. 2.21 shows the S-parameters of the balun. It works fine from 0.3 GHz to more than 18 GHz, with transmission losses of less than 1.9 dB over the entire band (in the frequencies of interest, up to 5.2 GHz, less than 0.8 dB). Note also that the hybrid mode is sufficiently attenuated, more than 20 dB, so that it does not degrade the behavior of the balun.. (a) QTEM Mode. (b) Hybrid Mode. Fig. 2.20: Electric field in the parallel strips’ port, and the QTEM (a) and the evanescent hybrid (b) modes.. Fig. 2.21: Simulated S-parameters of the microstrip to parallel strips balun. Note how the hybrid mode is quite attenuated with respect to the QTEM mode..

(35) CHAPTER 2. RADIOFREQUENCY HARVESTER. 2.4.1. 35. Back-to-Back. The model presented in Fig. 2.19 is now ready to be manufactured and to be used in the antenna, but first, it would be desirable to measure its response in order to prevent possible deviations in the final prototype. However, note that the balun is not measurable in our laboratory since it is a parallel strip configuration referred to an impedance of 188 Ω. In situations of this kind where measurement is not possible, as for example in the case of microstrip-to-slotline transitions (we cannot easily measure the slotline), the typical approach is to return to the original (measurable) line, in a scheme known as back-to-back. Figure 2.22 depicts the idea applied to our balun. Once the parallel strips are reached, we are going back to the original microstrip line by mirroring the tapered structure. The losses in the back-to-back configuration are approximately double those of the balun. This is reflected in the transmission parameter (green line) of Fig. 2.23, which indicates that losses are inferior to 3.8 dB in the whole band.. Fig. 2.22: Back-to-back configuration of the UWB microstrip to parallel strip balun..

(36) CHAPTER 2. RADIOFREQUENCY HARVESTER. 36. Fig. 2.23: Simulated S-parameters of the microstrip to parallel strips back-to-back configuration.. 2.4.2. Balun-Antenna Connection. Once we have designed the antenna, the balun, and its back-to-back configuration, the last simulation step is to unite the entire RF chain. It is not worth trying to simulate the joint structure, since it implies long simulations times due to a rather strong meshing (278 million cells), the long extension of the balun (the signals take time to propagate), and a large bandwidth (0.3 - 18 GHz). Besides that, welding is very poorly modeled in CST. Based on the fact that the balun minimally modifies the radiation pattern of the Archimedean spiral antenna, which was corroborated in simulation, the smarter way to simulate the joint structure is to model the antenna and the balun as two ”black boxes” (via its S-parameters) and to use CST Design Studio to study the composed response. This is reported in Fig. 2.24. As can be seen in Fig. 2.25, the antenna is well-matched from 0.38 GHz to 18 GHz, showing the good performance of the balun. Note now that the joint structure (antenna + balun of Fig. 2.24a) is fed via a 50-Ω SMA connector, the unbalance caused by attaching the connector directly to the arms of the antenna being eliminated by maintaining the ultrawideband response of the antenna.. (a) ”Physical” connec-(b) Use of CST Design Studio to make the tion. connection.. Fig. 2.24: Balun-antenna connection..

(37) CHAPTER 2. RADIOFREQUENCY HARVESTER. 37. Fig. 2.25: Reflection coefficient of the joint structure (antenna + balun).. 2.5. Measurements. Figure 2.26 presents a prototype of the miniaturized Archimedean spiral antenna. The antenna of the picture was constructed making use of a LPKF ProtoMat S100 milling machine [16] on a low-cost FR4 substrate (εr = 4.7, tan δ = 0.014 @1 MHz). As previously mentioned, the dimensions of the antenna are 19.77 cm x 20.72 cm.. Fig. 2.26: Prototype of the miniaturized Archimedean spiral antenna.. 2.5.1. Reflection Coefficient Measurement. We characterize in the laboratory the performance of the Archimedean spiral antenna in two measurement steps. In the first of them, we extract the reflection coefficient of the antenna from an Agilent 8722ES network analyzer, in order to check its bandwidth. Subsequently, the power spectrum is acquired with a N9020A MXA signal analyzer in two different scenarios: inside and outside the laboratory. In Fig. 2.27, it is shown a comparison between the simulated and measured reflection coefficients of the Archimedean spiral. As the experimental results probe (red line in Fig. 2.27), the antenna is well-matched (< −10 dB) from 350 MHz to 16 GHz, presenting an ultrawideband behavior. Besides that, its return loss is superior to 8.5 dB in the entire frequency range shown (0.3 – 20 GHz). On the other hand, although the measured reflection coefficient is slightly higher than the.

(38) CHAPTER 2. RADIOFREQUENCY HARVESTER. 38. Fig. 2.27: Comparison of the simulated and measured reflection coefficients of the Archimedean spiral antenna (normalized to 188.5 Ω). simulated one, note how the lower and upper cutoff frequencies coincide almost perfectly, being their values 350 MHz and 16 GHz, respectively.. 2.5.2. Power Spectrum Measurement. In order to obtain a realistic measurement of the harvested power, the antenna is connected to a signal analyzer capable of integrating the power spectrum over a prefixed bandwidth. Fig. 2.28 shows the power spectrum acquired by the miniaturized Archimedean spiral antenna in two different scenarios: inside and outside the laboratory (our laboratory is located inside the Higher Technical School of Telecommunication Engineers, Technical University of Madrid, and the nearest base station (LTE-800, GSM-900, GSM-1800) is situated at a distance of 60 meters and at a height of 8 meters from the receiving Archimedean antenna). As it can be noticed in Fig. 2.28, there is no significant contribution of power in any of them from 5200 MHz onwards, where the -65 dBm noise level prevails. From this frequency value, only militar and satcom applications can be found, but the directivity of the antenna is so reduced that it does not allow us to acquire anything remarkable. Despite capturing noise in a large frequency range (5.2 GHz – 16 GHz), its contribution to the total power harvested is negligible, hence it is not worthy to try to raise the upper cutoff frequency of the antenna (although, in this particular case, it is something intrinsic to its inner radius and it does not hinder the design). As seen in Fig. 2.28, the measured power level of the spectrum peaks is generally higher outside the laboratory than inside the laboratory, due to the radio signal attenuation caused by the building itself (position of the room, walls, windows, etc. [18]). Most of the energy captured comes from cellular bands, reaching almost -10 dBm for the 830 and 930 MHz bands. However, the measured FM spectral peak is higher than expected (-25 dBm outside the laboratory), even though the antenna is not specially designed to cover FM band. On the other hand, power contribution.

(39) CHAPTER 2. RADIOFREQUENCY HARVESTER. 39. Fig. 2.28: Power spectrum measured by the Archimedean spiral antenna inside and outside the laboratory with the most relevant bands remarked: 1-FM, 2-DTT, 3LTE-800, 4-GSM-900, 5-GSM-1800, 6-LTE-2100, 7-WiFi, 8-LTE-2600, 9-WiFi. of WiFi bands is surprisingly smaller than expected, leaving them in the background. Furthermore, it could be of interest comparing the power harvested by the Archimedean spiral antenna with other type of baseline antennas, such as dipoles or microstrip patches. In these terms, three different patches that operate within the 2.4 GHz band were designed and manufactured, as shown in Fig. 2.29. One of them (the one in the center of the image that is matched with a λ/4 transformer) is circularly polarized due to the slot inserted in the center of the patch, while the other two are linearly polarized. Fig. 2.30 shows a comparison between the power spectrum measured by the Archimedean spiral antenna (inside the laboratory) and by the three 2.4-GHz patches. The limited bandwidth of the microstrip patches, even within the WiFi band, naturally conditions the measured spectrum peaks to be smaller than in the case of using the Archimedean spiral antenna. The total harvested power, which is the available power between the arms of the Archimedean spiral antenna, is calculated as the sum of all carriers present in the radio spectrum. The result of integrating the power spectrum shown in Figs. 2.28 and 2.30 leads to the values presented in Table 2.1. We should remark that all measurements have been carried out in a realistic scenario, that is, where no directional sources have been intentionally put into scene. As previously commented, the power acquired outside the laboratory (1.86 dBm) is higher than the one acquired inside the laboratory (-3.19 dBm), both of which are sufficient to provide the power required (1–30 µW) by the elements discussed in Chapter 1..

(40) CHAPTER 2. RADIOFREQUENCY HARVESTER. 40. Fig. 2.29: Measured reflection coefficient of the manufactured 2.4-GHz patches (Dimensions: small patch: 4.89 x 3.91 cm, circularly-polarized patch: 5.05 x 3.91 cm, big patch: 5.02 x 4.07 cm.. Fig. 2.30: Comparison between the power spectrum measured by the Archimedean spiral antenna inside the laboratory and by the three 2.4-GHz patches, with the most relevant bands remarked: 1-FM, 2-DTT, 3-LTE-800, 4-GSM-900, 5-GSM-1800, 6LTE-2100, 7-WiFi, 8-LTE-2600, 9-WiFi..

(41) CHAPTER 2. RADIOFREQUENCY HARVESTER. 41. The best way to compare the amount of harvested power among antennas with different sizes is to express the acquired power per area unit, that is, the physical dimensions of the Archimedes spiral are 19.77 cm x 20.72 cm, which leads to 3.75 µW/cm2 and 1.17µW/cm2 outside and inside the laboratory, respectively. In Table 2.1, it is also presented the total power harvested by the three microstrip patches and an ultrawideband horn used in [30]. Note that despite the fact that the patch area is much smaller than the spiral area, the harvested power per area unit inside the laboratory is about ten times lower than in the case of the Archimedes spiral. On the other hand, it is also mentioned in the table that some measurements in [30] show that a power level of -10 dBm was acquired by a broadband horn, the nearest cell station being situated in this case at 150 meters. Table 2.1: Measurements on the harvested power. INSIDE THE LAB. Power Power/Area (dBm) (µW/cm2 ) Archimedean Spiral (0.3 – 16 GHz) Big Patch (2.4 GHz) Small Patch (2.4 GHz) Circularly- Polarized Patch (2.4 GHz) Horn (0.8-18 GHz) [30]. OUSIDE THE LAB. Power Power/Area (dBm) (µW/cm2 ). -3.19. 1.17. 1.86. 3.75. -26.01. 0.12. -. -. -28.68. 0.071. -. -. -26.69. 0.11. -. -. -10 (max). -. -. -.

(42) Chapter 3 Conditioning Circuit. Cockcroft-Walton Multiplier 3.1. Design of the Half-Wave Cockcroft-Walton Multiplier. A single stage of the half-wave Cockcroft-Walton (HWCW) multiplier, which is also called a Greinacher voltage doubler, is basically formed by two subcircuits [32]: a Villard circuit (circled in orange in Fig. 3.1), which is a diode clamping circuit, that is, a circuit capable of shifting the DC value of the input waveform; and a half-wave rectifier (circled in red in Fig. 3.1) that retains the peak value of the shifted signal. Essentially, the first capacitor charges to the peak AC voltage (Vpk ) on the negative half cycles, so that the output of the Villard circuit is the sum of the input waveform and the steady DC voltage of the capacitor in question, being ideally elevated to twice the voltage peak value (2Vpk ) on the positive half cycles, and ideally clamped to zero on its negative half cycles. Subsequently, the half-wave rectifier mitigates the enormous voltage ripple of the Villard circuit (2Vpk ) and smooths the clamped AC signal.. Fig. 3.1: Circuital scheme of the n-stage half-wave Cockcroft-Walton voltage multiplier: the Villard circuit and the half-wave rectifier are circled in orange and in red, respectively, and the Greinacher voltage doubler is remarked in dashed blue line.. 42.

(43) CHAPTER 3. CONDITIONING CIRCUIT. COCKCROFT-WALTON MULTIPLIER43 Cascading several stages allows us to multiply the output DC voltage value Vo in a proportional way to the number of stages n placed, that is Vo = 2nVpk − 2δV − ∆V,. (3.1). where 2δV is the output voltage ripple and 2∆V is a term that quantifies the voltage drop due to the incomplete charge of all capacitors when a load is connected (ideal components are assumed). In a no-load situation, charge and discharge times of the capacitors are nominally zero, so expression (3.1) can be simplified to Vo = 2nVpk . As depicted in [33], in a n-stage HWCW multiplier where all capacitors are identical (Fig. 3.1), the peak-to-peak voltage ripple is calculated as Il n(n + 1) q n(n + 1) = , (3.2) 2δV = C 2 fC 2 where q is the storage charge of all capacitors, f is the frequency of the input waveform, and Il is the load current. On the other hand, the voltage drop ∆V is calculated taking into account the amount of charge q that loses every stage at every cycle, that is, ∆V1 = Cq n, ∆V2 = Cq (2n + (n − 1)),..., ∆Vn = Cq (2n + 2(n − 1) + ... + 2 · 2 + 1). The sum of all contributions leads to q 2 3 1 2 1 Il 2 3 1 2 1 ∆V = n + n − n = n + n − n . (3.3) C 3 2 6 fC 3 2 6 Note that the first capacitors are most responsible of the voltage drop ∆V , as in the case of the voltage ripple 2δV , according to the progression ∆V1 , ∆V2 ,...,∆Vn mentioned above. Therefore, whether we increment their capacitances, both terms reduce and the output voltage Vo rises according to equation (3.1). However, as well mentioned in [32], it is only convenient doubling the first capacitor, since this one just has to face half of the voltage than the rest of capacitors. In that situation, ∆V1 is decreased by 0.5 Cq n, which reduces the voltage drop at every stage proportionally, leading to Il 2 3 1 n − n . (3.4) ∆VC1 =2C = fC 3 6 In a similar manner, it can be demonstrated that the voltage ripple is now of the form Il n2 2δVC1 =2C = . (3.5) fC 2 We have not considered yet the losses that the components present. In a first approximation, we can neglect the capacitor losses and focus only on the forward voltage drop of the diode, VF , which is very harmful in this particular case due to the low input voltages we are dealing with. As there are two diodes in every stage, there is a voltage drop of 2nVF due to the non-ideality of the diodes, being the output voltage Il 2 3 1 2 1 VoC1 =2C = 2n (Vpk − VF ) − n + n − n . (3.6) fC 3 2 6.

(44) CHAPTER 3. CONDITIONING CIRCUIT. COCKCROFT-WALTON MULTIPLIER44. 3.2. Measurements. The half-wave Cockcroft-Walton multiplier is a circuit commonly used in many areas of electronics, in particular in those areas related to the generation of high DC voltages (cathode ray tube television, particle accelerators, etc.), but when applied to energy harvesting in ultra-low power applications, there are two considerations that should be made. On the one hand, the input voltage peak value may be situated under 0.2 V in most cases, so it is crucial that the forward voltage drop in the diode is very low. On the other hand, the circuit operating frequency is about units of GHz, which requires a diode fast enough to follow the input signal acquired by the antenna. Two diodes that fulfill these two requirements are the HSMS-2822 (series mode), specially designed for input power levels above -20 dBm at frequencies below 4 GHz; and the HSMS-2850 (single mode), optimized for use in small signal (Pin < −20 dBm) applications at frequencies below 1.5 GHz. According to their datasheets [34,35], the two series diodes encapsulated in HSMS-2822 can provide 0.1 mA with a maximum voltage drop of 0.22 V (@ 25o C), while the HSMS-2850 diode can provide the same current with a maximum voltage drop of 0.15 V (@ 25o C).. (a) Top plane. (b) Bottom plane. Fig. 3.2: HWCW test set ((a) upper side, (b) bottom side) with six different combinations of components and number of stages: 33 nF – HSMS 2822 – 3 stages (blue), 33 nF// 33 pF – HSMS 2822 – 2 stages (purple), 33 pF – HSMS 2850 – 2 stages (green), 33 pF – HSMS 2822 – 2 stages (orange), 33 pF – HSMS 2850 (red) – 2 stages, 33 pF – HSMS 2822 – 5 stages (cyan)..

(45) CHAPTER 3. CONDITIONING CIRCUIT. COCKCROFT-WALTON MULTIPLIER45. Fig. 3.3: Measurement set-up: Agilent E4438C vector signal generator, Yokogawa DL9240L digital oscilloscope, HWCW test set, and a multimeter. In order to estimate the capacitor value to be used, a comparison between six different combinations of components and number of stages of the HWCW is carried out on the test board shown in Fig. 3.2. As well, Fig. 3.3 presents the measurement set-up used to characterize the response of each circuit. It is formed by Agilent E4438C vector signal generator [36], which operates from 250 kHz to 6 GHz; by Yokogawa DL9240L oscilloscope [37], a 1.5 GHz, 4-channel digital oscilloscope; and by a multimeter. Despite the quality of the oscilloscope, the probes available in the laboratory can only measure up to 500 MHz. Figure 3.4 presents the measured output DC voltage for different combinations of components in a 2-stage HWCW multiplier (no load connected). The input power of the sinusoidal waveform was set on -10 dBm, which generates an input voltage peak of 210 mV (measured with the digital oscilloscope). In a 2-stage ideal scheme, the output voltage should be 2 · 2 · 210 mV = 840 mV, but note that the output voltage decays to 500 mV in most schemes due to the voltage drop VF in the diodes. It is also observed a decay of the output voltage along the frequency, product of the parasitic elements associated to the capacitors and to the parasitics of the board. As can be noticed, the scheme that offers worst results is the one that uses the 33 nF capacitor (orange line), since it has its self-resonant frequency much lower (30 MHz) than the 33 pF capacitor (2.1 GHz). To palliate this effect and increase its range of operation, we added a parallel 33 pF capacitor (purple line), so that the equivalent capacitance is the same (33 nF + 33 pF ≈ 33 nF) but the self-resonant frequency of the equivalent capacitor is increased. In any case, there is no significant improvement. Using the same package (HSMS 2822), we obtain much better results placing the 33 pF capacitor (blue line), but in return, when a load is connected it shows a higher voltage ripple (20 mV @ 100 MHz). The circuit that offers better results is the one that uses the HSMS 2850 diode and.

(46) CHAPTER 3. CONDITIONING CIRCUIT. COCKCROFT-WALTON MULTIPLIER46. Fig. 3.4: Output DC voltages in a 2-stage HWCW scheme when using different components (no-load situation). the 33 pF capacitor (green line), even though the datasheet [35] indicates that the upper operating frequency is 1.5 GHz and that the recommended maximum input power is -20 dBm. However, when a load was connected to the circuit, the behavior of the diodes was degraded, so after removing the load and measuring the output voltage in a no-load situation, the output voltage was inferior to the measured before. Therefore, we do not recommend its use with an input power of -10 dBm. On the other hand, Fig. 3.5 shows a comparison of the output voltage versus frequency as a function of the input power (-10, -20 and – 30 dBm) on the 2-stage HWCW scheme, where HSMS 2822 diode and 33 pF capacitors are used. As expected, the lower the input power, the higher the percentage of power, in proportion, consumed by the diodes. This leads to a reduction of the output voltage as represented in the red line of Fig. 3.5, where for an input power of -30 dBm (1 µW) the circuit is not capable of raising the voltage properly (< 35 mV). Finally, Fig. 3.6 presents the output voltage, at each stage, of the 5-stage HWCW multiplier. It follows from Fig. 3.5 and Fig. 3.6 that it is not worthwhile placing several stages when the input power is so reduced. The intrinsic consumption of the diodes causes that the output power decreases non-linearly as a function of the input power. In summary, once the power transfer (from the antenna to the multiplier circuit) is optimized via the matching circuit, we must choose the appropriate number of stages that fulfills the output voltage requirement of the sensor (load). Nevertheless, note that the more stages are set, the higher the output voltage but the lower the output power. So, there is a trade-off between elevating the output voltage and maximizing the output power, what we are mainly interested in..

(47) CHAPTER 3. CONDITIONING CIRCUIT. COCKCROFT-WALTON MULTIPLIER47. Fig. 3.5: Output voltage versus frequency in the 2-stage HWCW scheme (with C=33 pF) at different input powers.. Fig. 3.6: Output voltage at each state of the 5-stage HWCW scheme (with C=33 pF) for an input power of -10 dBm..

(48) Chapter 4 Matching Circuit The analysis of the problem, from the point of view of the matching circuit, is depicted in Fig. 4.1. The source Zs and load ZL impedances of the two different matching circuit that maximize the power transfer from the RF harvester to the multiplier circuit initially depends on three parameters: the impedance of the antenna Zant , the number of stages N of the HWCW multiplier circuit, and the impedance of the load ZL placed. The non-linearity of the Cockcroft-Walton multiplier adds two other terms: the input power Pin , and the operating frequency f ; which greatly increase the complexity of the problem as seen in Fig. 4.1. Note that the optimum source impedance in the matching circuit is generally different from the antenna impedance. For the particular case of the Archimedean spiral antenna (Rant = 188 Ω), the measurements on the harvested power from the RF spectrum allow us to eliminate the power and frequency variables. Hence, the complexity of the problem is substantially reduced, as the source impedance of the matching circuit that ensures maximum power transfer is now only dependent on two terms: the number of stages N of the Cockcroft-Walton multiplier, and the selected load (the device to be fed).. Fig. 4.1: General scheme of the RF harvesting system. Note as the source impedance Zs depends on five terms: the input power Pin , the input range of frequencies f , the number of stages N in the Cockcroft-Walton multiplier , the intrinsic impendance of the antenna Zant , and the load ZL . 48.

(49) CHAPTER 4. MATCHING CIRCUIT. 49. Fig. 4.2: Scheme of the RF harvesting system particularized to the use of the Archimedean spiral antenna. Notes: *Pin1 = −5.80 dBm and Pin1 = −1.84 dBm, inside and outside the laboratory, respectively; and Pin2 = −9.20 dBm and Pin1 = −2.34 dBm, inside and outside the laboratory, respectively. **fin1 = 807 MHz, and fin2 = 942 MHz. Fortunately, the problem can be also divided into N sub-problems, which are only dependent on the load impedance, considering separately the different number of stages of the multiplier circuit.. 4.1. Narrow-band modeling of the Archimedean spiral antenna. We already know the impedance of the Archimedean spiral antenna is theoretically set to be 188 Ω, so the radiofrequency harvester can be modeled by a generator with and source impedance of Zant = 188 Ω. The generator, in turn, must contain the information relative to the power spectrum acquired by the antenna. It is not easy to model the spectrum power, due to the different carrier waveforms present in the spectrum and the multiple frequencies that are being taken into account (ultrawideband system). However, we do known that most part of the incoming power comes from 800/900-MHz bands, as can be noticed in Fig. 2.28. As Fig. 4.3 shows, both 800 and 900 MHz carriers are far for resembling to delta functions (sinusoidal excitations). However, modeling the carriers as two delta functions, placed at 807 MHz and 942 MHz, greatly simplifies the problem. The power peaks of both delta functions are calculated by integrating the power of both carriers along their range of frequencies, that is, integrating the power spectrum from 770 to 820 MHz and from 930 to 954 MHz, respectively. Fig. 4.3 depicts the process more clearly..

(50) CHAPTER 4. MATCHING CIRCUIT. 50. Fig. 4.3: Zoom over the power spectrum measured by the Archimedean spiral antenna at 800 and 900 MHz. The delta functions marked in blue and orange represent the modeling of the Archimedean spiral antenna as a generator, inside and outside the laboratory, respectively. In short, the Archimedean spiral antenna can be modeled from a circuital viewpoint as described in Fig. 4.4, the four parameters of interest being Pin1 , f1 = 807 MHz, Pin2 , and f2 = 942 MHz. Note that two different scenarios have been taken into account: the antenna being placed inside, and outside the laboratory.   −5.80 dBm if IN SIDE Pin1 = (4.1)  −1.84 dBm if OU T SIDE   −9.20 dBm if IN SIDE Pin2 = (4.2)  −2.34 dBm if OU T SIDE. Fig. 4.4: Narrow-band circuital model of the antenna..